Domain-distributed cryogenic signaling amplifier

ABSTRACT

A signal amplifier is distributed between first and second IC devices and includes a low-power input stage disposed within the first IC device, a bias-current source disposed within the second IC device and an output stage disposed within the second IC device. The output stage includes a resistance disposed within the second IC device and having a first terminal coupled to a drain terminal of a transistor within the input stage via a first signaling line that extends between the first and second IC devices.

CROSS REFERENCE TO RELATED APPLICATIONS

This application is a continuation of U.S. patent application Ser. No. 16/043,754 filed Jul. 24, 2018 (now U.S. Pat. No. 10,511,276), which claims priority to U.S. provisional application No. 62/536,456 filed Jul. 24, 2017. Each of the foregoing patent applications is hereby incorporated by reference.

TECHNICAL FIELD

The present disclosure relates to signaling between integrated circuit devices.

DRAWINGS

The various embodiments disclosed herein are illustrated by way of example, and not by way of limitation, in the figures of the accompanying drawings and in which like reference numerals refer to similar elements and in which:

FIG. 1 illustrates an embodiment of a chip-to-chip signaling system in which a multi-conductor signaling link is coupled between IC devices disposed in relatively cold and warm temperature domains;

FIG. 2 illustrates a more detailed view of input-stage, common-source bias and output-drive components of an AC-coupled complementary metal oxide semiconductor (complementary MOS or CMOS) amplifier and the distribution of those components within disparate cryogenic temperature domains on opposite sides of a signaling link;

FIG. 3 illustrates an alternative domain-distributed amplifier arrangement in which an RSFQ-sourced signal is coupled directly (instead of through capacitive elements) to gates of input-stage transistors;

FIG. 4 illustrates an alternative embodiment of an AC-coupled domain-distributed amplifier implemented with p-MOS transistors instead of n-MOS devices; and

FIG. 5 illustrates an embodiment of a double-data-rate (DDR) sampling circuit coupled to receive the output of a domain-distributed signal amplifier.

DETAILED DESCRIPTION

In various embodiments herein, signaling amplifier components are distributed between source and destination ICs (integrated circuit devices or chips) in different cryogenic temperature domains. In a number of embodiments, a low-power input stage component of an amplifier is implemented in a transmitting (signal source) IC disposed in a cryogenic temperature domain sufficiently cold to enable superconduction through Josephson junction stacks and/or operation of rapid single-flux quantum (RSFQ) circuits, while higher power biasing and output stage components of that same amplifier are implemented in a receiving (signal destination) IC device disposed in a substantially warmer (though possibly still cryogenic) temperature domain (cryogenic temperatures being, for example, temperatures below 93.15 K per U.S. National Institute of Standards and Technology, or, as occasionally defined, temperatures below 123 K). Although the source and destination ICs may implement virtually any core function, in particular embodiments the source IC implements a memory control function as in the case of a dedicated memory controller component or processor component (e.g., CPU) having a memory control function, while the destination IC implements a data storage function and/or signal buffering function as in the case of a memory component or buffer component, where the buffer component serves as an intermediary between the memory control/CPU component and one or more memory components. Although this source/destination terminology is carried forward in detailed embodiments presented below, in all cases, the destination IC may also transmit signals to the source IC, including transmitting signals to the source IC over the same signaling link(s) driven by the source IC (i.e., bidirectionally-driven signaling link(s)), and/or transmitting signals to the source IC via one or more dedicated (unidirectionally-driven) signaling link(s). For example, a source memory-control IC (e.g., CPU or dedicated memory controller) that transmits write data and/or control signals to a destination memory IC may also receive read data signals and/or status information transmitted by the memory IC

FIG. 1 illustrates an embodiment of a chip-to-chip signaling system 100 in which multi-conductor signaling link 110 is coupled between IC devices 101 and 111 disposed in relatively cold and warm temperature domains, respectively. More specifically, the cold-domain IC or “cold chip” 101 resides in a cryogenic domain cold enough to permit operation of superconducting circuit components (e.g., 4 Kelvin (4 K), though lower or higher temperatures may apply in other embodiments), while the warm-domain IC 111 (“warm chip”) resides in a warmer, but still cryogenic, 77 K environment (again, lower or higher temperatures may apply). As shown, cold chip 101 includes core circuitry 103 (i.e., to implement the core functions of the component, be they memory control, processing, etc.) together with physical signaling circuitry (PHY 105), both of which may be implemented predominantly by extremely low-power RSFQ circuit elements—an implementation that avoids the energy inefficiencies attendant in more conventional transistor-based circuitry. Warm chip 111 also includes core functional circuitry 113 (memory cell array, buffer, etc.) together with a PHY 115, but in a predominantly transistor-based implementation—for example in CMOS (complementary metal oxide semiconductor) circuitry.

While each of the PHYs within ICs 101 and 111 generally includes multiple transmitters and receivers (and/or transceivers) coupled to respective signaling links, PHYs 105 and 115 are depicted for purposes of explanation as minimally containing a synchronous differential transmitter and counterpart synchronous differential receiver, respectively. More specifically, on the transmit side (i.e., within cold chip 101), an RSFQ register element 108 is clocked by a transmit timing signal, CK_(T) (e.g., a “transmit clock” though a strobe signal may be employed instead), to deliver a stream of transmit data values to differential output driver 107 during a corresponding sequence of bit-time intervals (also referred to herein as unit intervals or bit intervals). More specifically, in one embodiment, the output of the RSFQ register is either a pulse or absence of a pulse within the corresponding bit interval (i.e., conveying logic ‘1’ and ‘0’ bit values, respectively, or logic ‘0’ and ‘1’ bit values) which, due to the quantum nature of the pulse (i.e., a ˜2.1 mV-picosecond pulse—h/2e, a magnetic flux quantum), is split into two or more identical outputs to drive respective driver branches within differential output driver 107. Output driver 107 responds to the incoming quantum pulse stream by driving, during each bit interval, a differential voltage or pseudo-differential voltage D±onto the component conductors of signaling link 110 with a voltage swing (difference between more positive and more negative voltage levels) and duration sufficient to enable data bit recovery within transistor-based PHY 115. In the depicted embodiment, receive-side PHY 115 includes transistor signal amplifier 117 (CMOS amplifier, though not limited to such) that amplifies the incoming differential signal (yielding amplified signal DX_(AMP)) to be sampled and latched within sampling element 118 in response to respective transitions of a receive timing signal, CK_(R). Note that while multi-conductor signaling link 110 is occasionally referred to herein as a differential signaling link, such terminology is intended to encompass conveyance of differential and pseudo-differential signals thereon.

When output driver 107 is implemented entirely by RSFQ circuits as shown at 130, disparately configured stacks of Josephson junctions 131, 133 (“driver branches”) respond to respective instances of the incoming transmit data value, DX_(SFQ)− (i.e., presence or absence of a quantum pulse), by generating a relatively high or relatively low voltage (or vice-versa) at a respective one of transmitter output nodes D+ and D−. More specifically, if the transmit data value (DX_(SPQ)−) supplied by register 108 is a logic ‘1’ as signified in complementary form by absence of a quantum pulse for the subject bit interval, driver branch 131 drives a relatively high voltage level at D+ while driver branch 133 drives a relatively low voltage level at D−, thus transmitting a logic ‘1’ voltage differential onto the differential signaling link—a transmission line in this case perceived by the transmitter as having an R_(O) (or Z_(O)) impedance on each conductor. Conversely, if the transmit data value is a logic 0, signified in complementary form by presence of a quantum pulse during the subject bit interval, D+ driver branch 131 drives a relatively low voltage level while D− driver branch 133 drives a relatively high voltage level to transmit a logic ‘0’ voltage differential onto signaling link 110. In either case, the resulting differential data eye (i.e., information bearing signal having an “eye width” duration and “eye height” differential amplitude) propagates across signaling link 110 to arrive at input nodes of CMOS signal amplifier 117.

While the RSFQ transmitter and receiver-side amplifier may be sufficient in some signaling applications, the relatively low signal levels produced at the outputs of driver branches 131 and 133 (and/or complexity of implementing the depicted Josephson-junction stacks) may present signal integrity issues (e.g., high error rates) in noisier environments and/or applications requiring longer signaling distances. On the other hand, simply moving CMOS amplifier 117 to the transmit-side 4K domain inflicts a substantial penalty as the power required to dissipate amplifier thermal losses is approximately 20× the amplifier power consumption—roughly 4 mW per signaling link in the case of a ˜200 uW CMOS amplifier, with such links numbering in the tens, hundreds, thousands or more in high-bandwidth multi-core quantum computing applications.

Observing that CMOS amplifier 117 includes a relatively low-power input stage component and relatively high-power bias and output-drive components, in a number of embodiments detailed below, the signal amplifier is split between the two cryogenic temperature domains, implementing the relatively low-power input stage component within the 4K domain as shown at 141, while leaving the relatively high-power bias component and output-drive component in the warmer 77K domain as shown at 143. Through this arrangement, a CMOS-amplified signal, DX_(T)±, is driven onto the signaling link by the 4K-domain circuitry, while the predominant share of total amplifier power dissipation occurs in the warmer 77K-domain circuitry. In the example shown, for instance, approximately 90% of the amplifier power is dissipated in the 77K domain, versus 10% power dissipation in the 4K domain—an arrangement that reduces the 4K-domain power penalty by a factor of 10 relative to wholesale relocation of the CMOS amplifier to the colder domain.

FIG. 2 illustrates a more detailed view of input-stage, common-source bias and output-drive components of an AC-coupled CMOS amplifier and the distribution of those components within disparate cryogenic temperature domains on opposite sides of a signaling link. Referring first to a single-domain amplifier implementation shown at 160, an incoming small-swing differential signal is ac-coupled (via capacitive elements 171, 172) to an amplifier input stage formed by gates of n-MOS transistors 175 and 176. Resistive elements R_(O), also forming part of the amplifier input stage, are coupled between a gate bias voltage source V_(GG) and respective gates of transistors 175 and 176 and serve both to bias the gates of transistors 175 and 176 at a desired DC-bias point (V_(GG)) and also to terminate the differential signaling link (i.e., nominally matching the link impedance). Current-source 177 is coupled between the common (interconnected) source terminals of transistors 175, 176 and ground and effectively establishes the transistor bias point, raising the source voltage (V_(S)) of each transistor to a level (and thereby establishing a gate-to-source bias voltage V_(GS)) that, in the absence of a differential input signal, will yield a balanced drain-to-source current (I_(DS)) of I_(O)/2 through each of transistors 175 and 176. Pull-up resistors R_(L) are coupled between respective drains of transistors 175 and 176—the differential output nodes of the amplifier—and supply voltage, V_(DD). When the incoming DX± signal components diverge to differential levels (i.e., to convey a data value), the high-going signal component will increase the small-signal gate-to-source voltage within the corresponding one of transistors 175 or 176, while the low-going component will produce the opposite effect (current decrease) within the other of those transistors, resulting in increased I_(DS) and thus reduced output voltage (due to the increased voltage drop across the corresponding R_(L) pull-up resistor) in one of the differential output nodes, and reduced I_(DS)/increased output voltage in the other of the output nodes. The bias points established by the V_(GG) potential and common-source bias current (I_(O)), and the amplifier gain established by the bias point and pull-up resistance values may vary according to application needs, but are generally chosen to yield a signal level that enables downstream CMOS sample and latch operation within a constrained power budget.

Still referring to FIG. 2, by relocating the relatively low power input stage components to the transmit-side of signaling link (4K domain), while leaving the relatively higher power transistor-bias current source and output-drive stage (pull-up) resistors in the 77K receive-side of the link, transmit-side signal amplification is achieved while maintaining the predominant amplifier power dissipation within the warmer 77K domain. In the specific example shown, pull-up resistors 181 and 182 (which now serve as link termination elements and are thus set to R_(O)) are set to approximately 50 ohms, V_(GG) is set to approximately one volt, V_(DD) is set to approximately 100 mV and transistor/common-source bias current I_(O) is set to approximately 2 mA—set points at which approximately 50 mV is dropped across each pull-up resistor (50 uW of power each), 10 mV is dropped across each of transistors 183 and 184 (10 uW of power each) and 40 mV is dropped across current source 185 (80 uW). Thus, of the ˜200 uW dissipated within the amplifier, only 20 uW (10%) is dissipated in the 4K domain, while 180 uW (90%) is dissipated within the warmer 77K domain, a vastly improved thermal result as compared with full amplifier instantiation within the 4K domain while retaining the signaling benefits of transmit-side amplification. Note that the bias points established by the bias-current source and gate bias voltage, and/or the amplification level established by the V_(DD) setpoint (in relation to the transistor bias point and the link-matched pull-up resistance elements) may vary in alternative embodiments through programmable adjustment (e.g., programmable current source) or design change. AC-coupling capacitors 187 and 188 may be sized similarly to those depicted at 171, 172 with implementation change as necessary to accommodate operation at 4 K.

FIG. 3 illustrates an alternative domain-distributed amplifier arrangement in which the RSFQ-sourced signal is coupled directly (instead of through capacitive elements 187, 188 as shown in FIG. 2) to gates of input-stage transistors 183 and 184. In this case, the gate bias voltage is lowered to the common-mode potential of the RSFQ output—from one volt to zero volts in the specific example shown—and the V_(DD) and V_(SS) voltage potentials are correspondingly lowered (i.e., from 100 mV to −900 mV and from 0V to −1V, respectively). The resistance values and current source setting are unchanged from those shown in the AC-coupled embodiment of FIG. 2, so that the voltage drops across transistors 183 and 184, current-source 185 and pull-up/termination resistances 181 and 182 are also unchanged. Consequently, the power dissipation split remains at 10% 4K domain and 90% 77K domain as desired.

FIG. 4 illustrates an alternative embodiment of an AC-coupled domain-distributed amplifier implemented with p-MOS transistors 203 and 204 instead of n-MOS devices and thus with transistor sources coupled in common to V_(DD) via biasing current source 205, output-stage/termination resistors 181, 182 being coupled between transistor drains and ground (effecting a pull-down configuration), and gate bias resistors R_(G) being coupled to a potential well below V_(DD) (i.e., V_(GG)=−800 mV in the depicted example) to establish the desired −V_(GS) bias point. V_(DD) and V_(SS) are set to to 100 mV and ground, respectively, and the bias current and pull-down resistance values (I_(O), R_(O)) match those in the embodiments shown in FIGS. 2 and 3 so that the power dissipation split remains, as in those cases, at 10% 4K, 90% 77K.

FIG. 5 illustrates an embodiment of a double-data-rate (DDR) sampling circuit 225 coupled to receive the output of a split-domain signal amplifier—in this instance, the AC-coupled split-domain amplifier depicted in FIG. 2. As shown, DDR sampler (which may be used to implement sampler 118 of FIG. 1) includes even-phase and odd-phase sample circuits 227 and 229, each coupled to receive the split-amplified differential data signal, DX_(AMP)±. Even-phase sampler 227 samples the incoming signal during even phases of the receive clock signal and thus in response to CK_(R)+, while odd-phase sampler 229 samples the input signal during odd phases of the receive clock signal (in response to CK_(R−)), each of samplers 227 and 229 latching one bit per clock cycle so that, as a whole, DDR sampler 225 captures (samples and latches) two bits per receive clock cycle.

In the FIG. 5 embodiment, even-phase and odd-phase samplers 227 and 229 are identically implemented by respective single-data-rate (SDR—capturing one bit per clock cycle) sample/latch circuits with the even-phase implementation shown in detail view 230. Transistors 241 and 243 (n-MOS devices) receive complementary components of the incoming differential signal at their respective gate terminals and are coupled in a common-source configuration with sample-enable transistor 245 (also n-MOS) coupled between the sources of transistors 241/243 and ground. Drain-interconnected p-MOS/n-MOS transistors 251/253 and 261/263 form respective inverters 250 and 260 coupled back-to-back (input of inverter 250 coupled to output of inverter 260 and vice-versa) to form a latch, with respective output nodes of those inverters (255, 265) forming the differential output of sampler 227. Precharge transistors 257 and 267 (p-MOS devices) are coupled source-to-drain between V_(DD) and the outputs of inverters 250 and 260, respectively, with gate terminals coupled to receive the incoming clock signal, CK_(R)+. Equalizing transistor 271 (also a p-MOS device) is coupled drain-to-source between the outputs of inverters 250 and 260 (and thus between their inputs as well) also with gate terminal coupled to receive CK_(R)+, and the gate terminal of sample-enable transistor 245 is coupled to receive CK_(R)+ as well. In this configuration, the low-phase of CK_(R)+ switches off sample-enable transistor 245, floating the sources of input-stage transistors 241/243 and thus the sources of n-MOS transistors 253 and 263 within respective inverters 250 and 260. The low-phase of CK_(R)+ also switches on precharge transistors 257 and 267 and equalizing transistor 271, precharging the inputs and outputs of inverters 250 and 260 to V_(DD) in preparation for a sampling operation during the ensuing high-phase of CK_(R)+. When CK_(R)+ goes high (switching on sample-enable transistor 245 and switching off transistors 257, 267 and 271), the differential input level at the gates of transistors 241 and 243 yields a relatively lower potential at the source of one of n-MOS transistors 241/243 than the other, powering on one of inverters 250, 260 before or more significantly the other so that the first-powered inverter (250 or 260) drives a low output in response to the precharged high input. Due to their back-to-back coupling, the low output of the first-powered inverter drives the input of other of inverters 250, 260 low to reinforce the high output from that other inverter and the low output from the first-powered inverter—snapping inverters 250 and 260 into a latched condition with one of inverters 250/260 driving a CMOS-high output and the other driving a CMOS-low output—a differential CMOS output signal—according to the state of the input differential signal, DX_(AMP)±. Odd-phase SDR sampler 229 operates in the same manner to precharge during the low phase of CK_(R)− (while a sample and latch operation is being carried out in SDR sampler 227) and to sample and latch the DX_(AMP)± signal during the high phase of CK_(R)− (while SDR sampler 227 is precharging).

It should be noted that the various circuits disclosed herein may be described using computer aided design tools and expressed (or represented), as data and/or instructions embodied in various computer-readable media, in terms of their behavioral, register transfer, logic component, transistor, layout geometries, and/or other characteristics. Formats of files and other objects in which such circuit expressions may be implemented include, but are not limited to, formats supporting behavioral languages such as C, Verilog, and VHDL, formats supporting register level description languages like RTL, and formats supporting geometry description languages such as GDSII, GDSIII, GDSIV, CIF, MEBES and any other suitable formats and languages. Computer-readable media in which such formatted data and/or instructions may be embodied include, but are not limited to, computer storage media in various forms (e.g., optical, magnetic or semiconductor storage media, whether independently distributed in that manner, or stored “in situ” in an operating system).

When received within a computer system via one or more computer-readable media, such data and/or instruction-based expressions of the above described circuits can be processed by a processing entity (e.g., one or more processors) within the computer system in conjunction with execution of one or more other computer programs including, without limitation, net-list generation programs, place and route programs and the like, to generate a representation or image of a physical manifestation of such circuits. Such representation or image can thereafter be used in device fabrication, for example, by enabling generation of one or more masks that are used to form various components of the circuits in a device fabrication process.

In the foregoing description and in the accompanying drawings, specific terminology and drawing symbols have been set forth to provide a thorough understanding of the disclosed embodiments. In some instances, the terminology and symbols may imply specific details that are not required to practice those embodiments. For example, any of the specific voltages, Josephson junction stack sizes or characteristic voltages/currents, signal path widths, signaling or operating frequencies, component circuits or devices and the like can be different from those described above in alternative embodiments. A signal driving circuit is said to “output” a signal to a signal receiving circuit when the signal driving circuit asserts (or de-asserts, if explicitly stated or indicated by context) the signal on a signal line coupled between the signal driving and signal receiving circuits. The term “coupled” is used herein to express a direct connection as well as a connection through one or more intervening circuits or structures. The terms “exemplary” and “embodiment” are used to express an example, not a preference or requirement. Also, the terms “may” and “can” are used interchangeably to denote optional (permissible) subject matter. The absence of either term should not be construed as meaning that a given feature or technique is required.

Various modifications and changes can be made to the embodiments presented herein without departing from the broader spirit and scope of the disclosure. For example, features or aspects of any of the embodiments can be applied in combination with any other of the embodiments or in place of counterpart features or aspects thereof. Accordingly, the specification and drawings are to be regarded in an illustrative rather than a restrictive sense. 

What is claimed is:
 1. A first integrated circuit die comprising: a bias voltage contact to receive a bias voltage from a second integrated circuit die; a bias current contact to receive a bias current generated by the second integrated circuit die; a first signal-transmit contact; and a first transistor to output, via the first signal-transmit contact, an information-bearing signal according to a digital input, the first transistor having a source terminal coupled to the bias current input such that at least part of the bias current flows into or out of the first transistor via the source terminal, a drain terminal coupled to the first signal-transmit contact, and a gate terminal coupled to the bias voltage contact and to receive the digital input.
 2. The first integrated circuit die of claim 1 further comprising logic circuitry to generate the digital input supplied to the gate terminal of the first transistor, the digital input constituting a data bit to be conveyed in the information-bearing signal and having either a first logic state or a second logic state.
 3. The first integrated circuit die of claim 2 wherein the logic circuitry to generate the digital input comprises circuitry to generate a pulse during a transmit interval to signal the first logic state of the data bit and to refrain from generating a pulse during the transmit interval to signal the second logic state of the data bit.
 4. The first integrated circuit die of claim 2 wherein the logic circuitry is capacitively coupled to the gate terminal of the first transistor.
 5. The first integrated circuit die of claim 2 wherein the logic circuitry comprises rapid single-flux quantum circuitry.
 6. The first integrated circuit die of claim 2 wherein the logic circuitry comprises one or more Josephson junctions.
 7. The first integrated circuit die of claim 1 further comprising a first resistive element and wherein the gate terminal of the first transistor is coupled to the bias voltage contact via the first resistive element.
 8. The first integrated circuit die of claim 1 further comprising a second signal-transmit contact and a second transistor to output, via the second signal-transmit contact, a complement of the information-bearing signal output via the first signal-transmit contact.
 9. The first integrated circuit die of claim 8 wherein the second transistor comprises a source terminal coupled to the bias current input, a drain terminal coupled to the second signal-transmit contact, and a gate terminal coupled to the bias voltage contact and to receive a complement of the digital input.
 10. The first integrated circuit die of claim 1 wherein the first transistor is a P-type metal oxide semiconductor (PMOS) transistor through which the bias current flows from source to drain such that the bias current flows into the first integrated circuit die via the bias current contact and flows out of the first integrated circuit die via the first signal-transmit contact.
 11. The first integrated circuit die of claim 1 wherein the first transistor is an N-type metal oxide semiconductor (NMOS) transistor through which the bias current flows from drain to source such that the bias current flows into the first integrated circuit die via the first signal-transmit contact and flows out of the first integrated circuit die via the bias current contact.
 12. A method of transmitting an information-bearing signal within a first integrated circuit die, the method comprising: receiving a bias voltage and bias current from a second integrated circuit die; biasing a first transistor during a transmit interval with the bias voltage and bias current, the first transistor having a drain terminal coupled to a signal-transmit contact of the first integrated circuit die; and supplying an input signal to a gate terminal of the first transistor during the transmit interval to produce, as the information bearing signal, an output signal at the signal-transmit contact, the output signal being amplified relative to the input signal according the biasing of the first transistor.
 13. The method of claim 12 wherein biasing the first transistor comprises establishing a voltage between gate and source terminals of the first transistor in accordance with the bias voltage and the bias current.
 14. The method of claim 12 wherein biasing the first transistor comprises applying the bias voltage to the gate terminal of the first transistor via a resistive element and conducting the bias current between the source and drain terminals of the first transistor.
 15. The method of claim 12 wherein the first transistor is a P-type metal oxide semiconductor (PMOS) transistor through which the bias current is conducted from the source terminal to the drain terminal, and wherein receiving the bias current from the second integrated circuit die comprises receiving a current sourced by the second integrated circuit die.
 16. The method of claim 12 wherein the first transistor is a N-type metal oxide semiconductor (NMOS) transistor through which the bias current is conducted from the drain terminal to the source terminal and wherein receiving the bias current from the second integrated circuit die comprises drawing the bias current from the source terminal of the first transistor to a current source within the second integrated circuit die.
 17. The method of claim 12 wherein supplying the input signal to the gate terminal of the first transistor comprises conveying the input signal to the gate terminal of the first transistor via a capacitive coupling.
 18. The method of claim 17 wherein supplying the input signal to the gate terminal of the first transistor during the transmit interval comprises supplying the input signal to the gate terminal in either a first logic state represented by a voltage pulse during the transmit interval or in a second logic state represented by absence of a voltage pulse during the transmit interval.
 19. The method of claim 17 wherein supplying the input signal to the gate terminal of the first transistor during the transmit interval comprises generating the input signal within circuitry having at least one of (i) rapid single-flux quantum circuitry or (ii) one or more Josephson junctions.
 20. A first integrated circuit die comprising: means for biasing a first transistor during a transmit interval with a bias voltage and bias current received from a second integrated circuit die, the first transistor having a drain terminal coupled to a signal-transmit contact of the first integrated circuit die; and means for supplying an input signal to a gate terminal of the first transistor during the transmit interval to transmit an information-bearing output signal at the signal-transmit contact, the output signal being amplified relative to the input signal according the biasing of the first transistor. 